1. Technical Field
The present invention relates to operational amplifiers (xe2x80x9cop-ampsxe2x80x9d) and more particularly to a rail-to-rail input stage of a CMOS op-amp having a constant transconductance which is independent of the common-mode input voltage.
2. Background Information
An exemplary two-stage op-amp configuration 10 is illustrated in FIG. 1. Op-amp 10 contains amplifier stage 100 and amplifier stage 102. Amplifier stage 100 comprises a transconductance amplifier with a differential input stage, i.e., there are two input terminals in amplifier stage 100: negative input 106 and positive input 108. Amplifier stage 100 is configured to provide an output current to amplifier stage 102 that is proportional to the difference in voltage between input 106 and 108.
Amplifier stage 102 comprises a high-gain amplifier. A capacitor 104 is connected in a feedback loop between an output 110 of amplifier stage 102 and an input 112 of amplifier stage 102. Capacitor 104 is present to ensure that the op-amp is stable when the op-amp is operated in a feedback configuration. For an amplifier stage 102 with a sufficiently large gain, the total gain of amplifier stage 100 and amplifier stage 102 is Gm/sC, where Gm is the transconductance of amplifier stage 100 and C is the capacitance of capacitor 104. Thus, the op-amp has the frequency response of a low-pass amplifier, as illustrated in FIG. 2. The gain versus frequency curve 200 shows that the gain is reasonably stable at low frequencies, but is continually reduced at higher frequencies. Corner frequency 210 is approximately the frequency at which the gain starts decreasing.
For operation, amplifiers require a power source. This power source is typically in the form of a supply voltage. While supply voltages in the range of 5 to 10 volts were largely used in the past, supply voltages have more recently decreased to below 3 volts, with supply voltages below 1 volt being introduced. At these low voltages, it is commonly desired for an op-amp to operate at input voltages close to that of the power supply to facilitate a larger range of operation. This operational characteristic is termed xe2x80x9crail-to-railxe2x80x9d operation.
An op-amp circuit using only P-type transistors can only operate within a voltage range from the negative supply rail to the positive supply rail minus the gate-source voltage, VGS, and the saturation voltage, Vdsat, of a tail current source. Analogously, an op-amp circuit using only N-type transistors can operate only from the positive supply rail down to VGS and Vdsat above the negative rail voltage. Accordingly, in order to achieve rail-to-rail operation, a circuit must use both P-type transistors and N-type transistors.
One circuit that illustrates a CMOS differential input stage of a rail-to-rail op-amp is shown in FIG. 3. The input stage comprises two pairs of input transistors driven in parallel: P-type transistors 300 and 302; and N-type transistors 304 and 306. A current source 308 supplies the current for P-type transistors 300 and 302 while a current source 310 supplies the current for N-type transistors 304 and 306. A negative terminal 320 and a positive terminal 322 are the input terminals for this differential amplifier. Both negative terminal 320 and positive terminal 322 are coupled to both an N-type transistor and a P-type transistor. Specifically, negative terminal 320 is coupled to P-type transistor 300 and to N-type transistor 304; positive terminal 322 is coupled to P-type transistor 302 and N-type transistor 306.
One problem with the circuit illustrated in FIG. 3 is the resulting change in the transconductance of the circuit. This problem can be illustrated in the graph of FIG. 4, where axis 410 represents the transconductance Gm of the circuit of FIG. 3 and axis 420 represents the common-mode input voltage.
In region 400, only the P-type transistors are operating such that the transconductance of the circuit comprises only the transconductance of the P-type transistors. In region 404, only the N-type transistors are operating such that the transconductance of the circuit comprises only the transconductance of the N-type transistors. Ideally, the circuit is constructed such that the transconductance of the N-type transistors is approximately the same as the transconductance of the P-type transistors. Therefore, the transconductance in region 400 is equal to the transconductance in region 404. However, in a region 402, wherein both pairs of transistors are operating, the transconductance of the circuit in region 402 comprises the sum of the transconductance of the N-type transistors and the transconductance of the P-type transistors. Because the transconductances for both types of transistors are ideally equal, the total transconductance in region 402 is approximately double the transconductance of the circuit in region 400 and region 404.
It is not desirable to have a transconductance that varies with the common-mode input voltage. As explained above, the gain of an op-amp using this type of configuration is linearly related to the transconductance of amplifier stage 100 (gain=Gm/sC). Since the gain of the op-amp is dependent on the transconductance Gm of amplifier stage 100, the gain of the op-amp is not constant. In addition, the frequency response of the op-amp varies if transconductance Gm is not constant, as the time constant of the circuit varies with Gm. Accordingly corner frequency 210 of FIG. 2 tends to vary, resulting in an unstable frequency response.
As described in Johan H. Huijsing et al., Low-Power Low-Voltage VLSI Operational Amplifier Cells, IEEE Transactions on Circuits and Systems, Vol. 42, No. 11 (November 1995), the problem described above is also present in circuits using bipolar transistors. One solution for bipolar circuits, according to Huijsing et al., is to keep constant the sum of the tail currents for the N-type transistors and for the P-type transistors.
An application of the Huijsing et al. solution to FET circuits is shown in FIG. 5. Transistors 300, 320, 304, and 306 are identical to those shown in FIG. 3. It should be noted that the connections from transistors 300, 320, 304, and 306 to the next stage are omitted to facilitate a discussion of FIG. 5. Current source 308 is analogous to current source 308 in FIG. 3. However, there is no separate current source for the N-type transistors. Additional transistors 526, 528, and 530, along with a voltage source 524, are configured to direct the current from current source 308 to supply the N-type transistors. Specifically, transistor 526 is a current transfer transistor while transistors 528 and 530 comprise a current mirror that supplies the current to the N-type transistors. Meanwhile, voltage source 524 biases transistor 526 such that transistor 526 is in a proper operating mode. Accordingly, the total supply current in the circuit is kept constant, i.e., the P-type transistors are directly supplied current by current source 308, while the N-type transistors are indirectly supplied current by current source 308 through use of transistors 526, 528, and 530.
At low input voltages, only P-type transistors 300 and 302 are operating, each being supplied current by current source 308 and generating output tail currents 550 and 552 at their respective drains. Although not shown, tail currents 550 and 552 may be summed and propagated to the next stage of the op-amp. At high input voltages, only N-type transistors 304 and 306 are operating. In this case, no current is being supplied to the P-type transistors. Current source 308 supplies current to the N-type transistors 304 and 306 though transistors 526, 528, and 530, with resulting output tail currents 554 and 556 being present at the drains of N-type transistors 304 and 306. Although not shown, tail currents 554 and 556 may also be summed and propagated to the next stage of the op-amp. Therefore, when an input pair, such as input transistors 300 and 302 or transistors 304 and 306, is operating, the input pair is being supplied current by a current source, with a non-zero tail current being present.
As discussed, in region 402 of FIG. 4, both the P-type input pair and the N-type input pair are operating. Thus, both input pairs are being supplied with current, e.g., the P-type input pair being directly supplied by current source 308 and the N-type input pair being supplied through transistors 526, 528, and 530.
However, the above configuration does not operate optimally if the FETs are not biased during weak inversion, i.e., when the gate is biased below the threshold voltage. Moreover, if the FETs are biased during strong inversion, i.e., when the gate voltage is larger than the threshold voltage, transconductance Gm still varies by approximately 40% since transconductance Gm is proportional to the square root of the drain current. In contrast, transconductance Gm is linearly proportional to the drain current for both BJTs and FETs during weak inversion.
The Huijsing et al. reference further suggests the use of an op-amp circuit that supplies each of the input transistor pairs with four times the normal tail current when the other pair is switched off. Huijsing et al. discloses that a transconductance Gm that varies by about 15% across the amplifier""s operating range can be realized. However, many applications today require the further reduction of the variation of transconductance Gm significantly below that available from the prior art.
The present invention addresses many of the shortcomings of the prior art. In accordance with one aspect of the present invention, an operational amplifier circuit comprising a differential input stage includes an input stage and a proportional input stage and a minimum selector circuit. In accordance with an exemplary embodiment, the minimum selector circuit is suitably configured to receive two input currents provided by the input stage and the proportional input stage, and then output the minimum current to a current transfer circuit. The current transfer circuit is suitably coupled to the input stage. The minimum current can be suitably subtracted from the output current of the current transfer circuit to reduce the total transconductance of the operational amplifier circuit.